viernes, 23 de julio de 2010
Power MOSFETs are well known for superior switching speed, and they require very little gate drive power because of the insulated gate. In these respects, power MOSFETs approach the characteristics of an "ideal switch". The main drawback is on-resistance RDS(on) and its strong positive temperature coefficient. This application note explains these and other main features of high voltage N-channel power MOSFETs, and provides useful information for device selection and application. Microsemi's Advanced Power Technology MOSFET datasheet information is also explained.
Power MOSFET structure
Figure 1 shows a cross section of an APT N-channel power MOSFET structure. (Only N-channel MOSFETs are discussed here.) A positive voltage applied from the source to gate terminals causes electrons to be drawn toward the gate terminal in the body region. If the gate-source voltage is at or above what is called the threshold voltage, enough electrons accumulate under the gate to cause an inversion n-type layer; forming a conductive channel across the body region (the MOSFET is enhanced). Electrons can flow in either direction through the channel. Positive (or forward) drain current flows into the drain as electrons move from the source toward the drain. Forward drain current is blocked once the channel is turned off, and drain-source voltage is supported by the reverse biased body-drain p-n junction. In N-channel MOSFETs, only electrons flow during forward conduction — there are no minority carriers. Switching speed is only limited by the rate that charge is supplied to or removed from capacitances in the MOSFET. Therefore switching can be very fast, resulting in low switching losses. This is what makes power MOSFETs so efficient at high switching frequency.
On Resistance RDS(on)
The main components of on-resistance RDS(on) include the channel, JFET (accumulation layer), drift region, and parasitics (metallization, bond wires, and package). At voltage ratings above about 150V, drift region resistance dominates RDS(on).
Figure 2 RDS(on) vs. Current
The effect of current on RDS(on) is relatively weak in high voltage MOSFETs . Looking at Figure 2, doubling the current results in only about a 6% increase in RRDS(on).
Figure 3 RDS(on) vs. Temperature
Temperature on the other hand has a strong effect on RRDS(on). As seen in Figure 3, on resistance approximately doubles from 25°C to 125°C. The temperature coefficient of RDS(on) is the slope of the curve in Figure 3 and is always positive because of majority-only carriers. The strong positive RDS(on) temperature coefficient compounds the I2R conduction loss as temperature increases.
The positive RDS(on) temperature coefficient is a nice feature when paralleling power MOSFETs because it ensures thermal stability. It does not however ensure even current sharing. This is a common misconception. What really makes MOSFETs so easy to parallel is their relatively narrow part-to-part parameter distribution, particularly RRDS(on), combined with the security from current hogging provided by the positive RDS(on) temperature coefficient.
For any given die size, RDS(on) also increases with increasing voltage rating V(BR)DSS, as shown Figure 4.
Figure 4 Normalized RDS(on) vs. V(BR)DSS
A curve fit of rated RDS(on) versus V(BR)DSS for Power MOS V and Power MOS 7 MOSFETs reveals that RRDS(on) increases as the square of V(BR)DSS. This nonlinear relationship between RDS(on) and V(BR)DSS is a compelling reason to research ways to reduce the conduction loss of power transistors.
Within the structure of a MOSFET, you can imagine an integral JFET shown in Figure 1. This JFET has a significant influence on RDS(on) and is part of the normal operation of the MOSFET.
Intrinsic body diode
The body-drain p-n junction forms an intrinsic diode called the body diode (see Figure 1). Reverse drain current cannot be blocked because the body is shorted to the source, providing a high current path through the body diode. Enhancing the device reduces conduction loss when reverse drain current flows because electrons flow through the channel in addition to electrons and minority carriers flowing through the body diode.
The intrinsic body diode is convenient in circuits that require a path for reverse drain current (often called freewheeling current), such as bridge circuits. For these circuits, FREDFETs are offered with improved reverse recovery characteristics. FREDFET is simply a trade name Advanced Power Technology uses to distinguish a MOSFET that has additional processing steps to speed up the reverse recovery of the intrinsic body diode. There is no separate diode in a FREDFET; it is the MOSFET intrinsic body diode. Either electron irradiation (which is usually used) or platinum doping is used for minority carrier lifetime control in the body diode, greatly reducing the reverse recovery charge and time.
A side effect of FREDFET processing is higher leakage current, particularly at high temperature. However, considering that MOSFETs have low leakage current to begin with, the added leakage current of a FREDFET is normally of no concern below 150°C junction temperature. Depending on the irradiation dose, a FREDFET may have a higher RDS(on) rating than a corresponding MOSFET. The body diode forward voltage is also slightly higher for a FREDFET. Gate charge and switching speed are identical between MOSFETs and FREDFETs. From here on, the term MOSFET will be used for both MOSFETs and FREDFETs unless specifically stated otherwise.
The reverse recovery performance of a MOSFET or even of a FREDFET is "crummy" compared to a discrete fast recovery diode. In a hard switched application operating at 125°C, the turn-on loss in the switch due to the reverse recovery current of the body diode is about five times higher than if a discrete fast recovery diode is used. There are two reasons for this:
1.-The area of the body diode is the same as the area of the MOSFET or FREDFET, whereas the area of a discrete diode for the same function can be much smaller and hence have much lower recovery charge.
2.- The body diode of a MOSFET or even a FREDFET is not optimized for reverse recovery like a discrete diode is.
Like any conventional silicon diode, body diode reverse recovery charge and time depend on temperature, di/dt, and current. The forward voltage of the body diode, VSD, decreases with temperature by about 2.5 mV/°C.
Parasitic bipolar transistor
The layered MOSFET structure also forms a parasitic NPN bipolar junction transistor (BJT), and turning it on is definitely not part of normal operation. If the BJT were to turn on and saturate, it would result in a condition called latchup, where the MOSFET cannot be turned off except by externally interrupting the drain current. High power dissipation during latchup can destroy the device.
The base of the parasitic BJT is shorted to the source to prevent latchup and because breakdown voltage would be greatly reduced (for the same RDS(on)) if the base were allowed to float. It is theoretically possible for extremely high dv/dt during turn-off to cause latchup. For modern, conventional power MOSFETs however, it is very difficult to build a circuit capable of achieving such high dv/dt.
There is a risk of turning on the parasitic BJT if the body diode conducts and then commutates off with excessively high dv/dt. High commutation dv/dt causes high current density of minority carriers (positive carriers, or holes) in the body region, which can build up enough voltage across the body resistance to turn on the parasitic BJT. This is the reason for the peak commutating (body diode recovery) dv/dt limit in the datasheet. Peak commutating dv/dt is higher for a FREDFET compared to a MOSFET because of reduced minority carrier lifetime.
Switching speed and loss are practically unaffected by temperature because the capacitances are unaffected by temperature. Reverse recovery current in a diode however increases with temperature, so temperature effects of an external diode (be it a discrete diode or a MOSFET or FREDFET body diode) in the power circuit affect turn-on switching loss.
The threshold voltage, denoted as VGS(th), is really a turn-off specification. It tells how many milliamps of drain current will flow at the threshold voltage, so the device is basically off but on the verge of turning on. The threshold voltage has a negative temperature coefficient, meaning the threshold voltage decreases with increasing temperature. This temperature coefficient affects turn-on and turn-off delay times and hence the dead-time requirement in a bridge circuit.
Figure 5 Transfer Characteristic Example
Figure 5 shows the transfer characteristic for an APT50M75B2LL MOSFET. The transfer characteristic depends on both temperature and drain current. In Figure 5, below 100 Amps the gate-source voltage has a negative temperature coefficient (less gate-source voltage at higher temperature for a given drain current). Above 100 Amps, the temperature coefficient is positive. The gate-source voltage temperature coefficient and the drain current at which it crosses over from negative to positive are important for linear mode operation.
Breakdown voltage has a positive temperature coefficient, as will be discussed in the Walkthrough section.
Short circuit capability
Short circuit withstand capability is not typically listed in the datasheet. This is simply because conventional power MOSFETs are unmatched for short circuit withstand capability compared to IGBTs or other devices with higher current density. It goes without saying that MOSFETs and FREDFETs are short circuit capable.
VDSS — Drain-source voltage
This is a rating of the maximum drain-source voltage without causing avalanche breakdown, with the gate shorted to the source and the device at 25°C. Depending on temperature, the avalanche breakdown voltage could actually be less than the VDSS rating. See the description of V(BR)DSS in Static Electrical Characteristics.
VGS — Gate-source voltage
VGS is a rating of the maximum voltage between the gate and source terminals. The purpose of this rating is to prevent damage of the gate oxide. The actual gate oxide withstand voltage is typically much higher than this but varies due to manufacturing processes, so staying within this rating ensures application reliability.
ID — Continuous drain current
ID is a rating of the maximum continuous DC current with the die at its maximum rated junction temperature TJ(max) and the case at 25°C and sometimes also at a higher temperature. It is based on the junction-to-case thermal resistance rating RθJC and the case temperature TC as follows:
This equation simply says that the maximum heat that can be dissipated,
equals the maximum allowable heat generated by conduction loss, I2D X RDS(on)@TJ(max), where RDS(on)@TJ (max) is the ON-resistance at the maximum junction temperature.
Solving for ID
Note that there are no switching losses involved in ID, and holding the case at 25°C is seldom feasible in practice. Because of this, actual switched current is typically less than half of the ID @ TC = 25°C rating in a hard switched application; one fourth to one third is common.
Graph of ID versus TC
This graph is simply the solution to (2) over a range of case temperatures. Switching losses are not included. Figure 6 shows an example. Note that in some cases, the package leads limit the continuous current (switched current can be higher): 100 Amps for TO-247 and TO-264 packages, 75 Amps for TO-220 package, and 220 Amps for the SOT-227 package.
Figure 6 Maximum drain current vs. case temperature
IDM — Pulsed drain current
This rating indicates how much pulsed current the device can handle, which is significantly higher than the rated continuous DC current. The purposes of the IDM rating are:
To keep the MOSFET operating in the Ohmic region of its output characteristic. See Figure 7. There is a maximum drain current for a corresponding gate-source voltage that a MOSFET will conduct. If the operating point at a given gate-source voltage goes above the Ohmic region "knee" in Figure 7, any further increase in drain current results in a significant rise in drain-source voltage (linear mode operation) and a consequent rise in conduction loss. If power dissipation is too high for too long the device may fail. The IDM rating is set below the "knee" for typical gate drive voltages.
A current density limit to prevent die heating that otherwise could result in a burnout site.
To avoid problems with excessive current through the bond wires in case the bond wires are the "weak link" instead of the die.
Figure 7 MOSFET Output Characteristic
PD — Total power dissipation
This is a rating of the maximum power that the device can dissipate and is based on the maximum junction temperature and the thermal resistance RqJC at a case temperature of 25°C.
The linear derating factor is simply the inverse of RθJC.
TJ, TSTG — Operating and storage junction temperature range
This is the range of permissible storage and operating junction temperatures. The limits of this range are set to ensure a minimum acceptable device service life. Operating well within the limits of this range can significantly enhance the service life.
EAS — Single pulse avalanche energy
If a voltage overshoot (typically due to leakage and stray inductances) does not exceed the breakdown voltage, then the device will not avalanche and hence does not need to dissipate avalanche energy. Avalanche energy rated devices offer a safety net for over-voltage transients, depending on the amount of energy dissipated in avalanche mode.
All devices that are avalanche energy rated have an EAS rating. Avalanche energy rated is synonymous with unclamped inductive switching (UIS) rated. EAS indicates how much reverse avalanche energy the device can safely absorb.
Conditions for a test circuit are stated in a footnote, and the EAS rating is equal to
where L is the value of an inductor carrying a peak current iD, which is suddenly diverted into the drain of the device under test. It is the inductor voltage exceeding the breakdown voltage of the MOSFET that causes the avalanche condition. An avalanche condition allows the inductor current to flow through the MOSFET, even though the MOSFET is in the off state. Energy stored in the inductor is analogous to energy stored in leakage and/or stray inductances and is dissipated in the MOSFET.
When MOSFETs are paralleled, it is highly unlikely that they have exactly the same breakdown voltage. Typically, one device will avalanche first and subsequently take all the avalanche current (energy).
EAR — Repetitive avalanche energylanche energy
A repetitive avalanche rating has become "industry standard" but is meaningless without information about the frequency, other losses, and the amount of cooling. Heat dissipation (cooling) often limits the repetitive avalanche energy. It is also difficult to predict how much energy is in an avalanche event. What the EAR rating really says is that the device can withstand repetitive avalanche without any frequency limitation, provided the device is not overheated, which is true of any avalanche capable device. During design qualification, it is good practice to measure the device or heat sink temperature during operation to see that the MOSFET does not overheat, especially if avalanching is possible.
IAR — Avalanche current
For some devices, the propensity for current crowding in the die during avalanche mandates a limit in avalanche current IAR. Thus avalanche current is the "fine print" of avalanche energy specifications; it reveals the true capability of a device.
Freddy Vallenilla. EES
Publicado por Tecnología en Telecomunicaciones - conocimientos.com.ve en 20:33
Etiquetas: 1II 2010-1 CAF Freddy Vallenilla